Adaptive filtering for more reliably determining physiological parameters

ABSTRACT

Apparatus for reducing an interference portion in a time-discrete signal further including a useful portion, including a first provider for providing the time-discrete signal including the interference portion and the useful portion; a second provider for providing a first time-discrete reference signal including a first interference portion, and a second time-discrete reference signal including a second interference portion, the second interference portion being shifted in phase relative to the first interference portion. The apparatus further includes a subtractor for generating a differential signal from the two reference signals, the differential signal including a frequency component caused by the first and second interference portions; and a manipulator for manipulating the time-discrete signal on the basis of the differential signal such that in a manipulated time-discrete signal the frequency component is reduced.

BACKGROUND OF THE INVENTION

The present invention relates to an apparatus for reducing an interference portion in a time-discrete signal for more reliably determining a physiological parameter from the time-discrete signal. The method may be applied in plethysmogram-based measuring methods (e.g. plethysmography, pulse oximetry) for the purpose of suppressing aliasing interferences.

Plethysmography is an optical method of obtaining a so-called plethysmogram which provides information about the pulse frequency and the oxygen saturation of the blood of a subject. A plethysmogram is a graphic illustration of volume changes. In this field of application, it is specifically the volume changes of an arterial blood flow at a localized measurement site of the human body which are detected as the plethysmogram. To implement this in technical terms, light is radiated through tissue at a body location having arterial blood vessels. The patient has a sensor applied to him/her which contains a light source and a photoreceiver, so that the light passes through the tissue layer, and so that the remaining light intensity impinges upon the photoreceiver. Within the body, the light undergoes an attenuation which is dependent, among other things, on the wavelength of the light source, the type and concentration of the substances in the irradiated tissue, and on the pulsation of the blood. The signal of the photoreceiver which has thus been obtained is present in the form of a photocurrent, is dependent on the above-mentioned general conditions, and corresponds, in a first approximation, to the changes in the blood volume of arterial vessels which are caused by contraction of the heart muscle. FIG. 24 shows the basic architecture of an apparatus for detecting a plethysmogram. A microcontroller (μC) controls, via two driver stages, two LEDs of different wavelengths, one light source being sufficient, in principle, for creating a plethysmogram. The LEDs depicted in FIG. 24 emit light in the red and infrared regions. The light emitted by the LEDs subsequently passes through the tissue of the subject, in FIG. 24 this is depicted, by way of example, as a finger. Once the light has passed through the tissue of the subject, it will impinge upon a photosensor. The photosensor converts the optical signals into electrical signals and passes them on to processing electronics which amplify the signal, convert them from analog to digital and feed them to the microcontroller (μC). The microcontroller (μC) then determines two plethysmograms, one plethysmogram for each wavelength, from the digital signals fed to it. From the waveforms of the plethysmograms thus measured, physiological parameters, such as the heart rate or the oxygen saturation of the subject's blood, may be measured, with one single plethysmogram being sufficient, in principle, for determining the heart rate, for determining the oxygen saturation of the blood, two plethysmograms of light sources of different wavelengths being useful.

Pulse oximetry is a non-invasive method of measuring the oxygen saturation of the blood (SpO₂) and the heart rate (HR) by means of an optical sensor. The oxygen saturation detected by the pulse oximeter is specifically referred to as the SpO₂ value. The oxygen saturation is defined as the ratio of the concentration of oxygen-saturated hemoglobin molecules and the overall hemoglobin concentration, and is indicated in percent. A component of the pulse oximeter is a sensor having two integrated light sources and being configured similar to a plethysmograph, cf. FIG. 24. In pulse oximetry, use is made of at least two plethysmograms to determine the color of the arterial blood. The color of the blood, in turn, is dependent on the oxygen saturation. By selecting the wavelengths of the light sources well, it may be shown that a quantity correlating well with the oxygen saturation may be obtained from the ratios of prominent points within the plethysmogram. Typically, the spectra of the receive signals of two light sources of different wavelengths are determined, and the quotient of specific spectral values is formed. This quotient will then be approximately proportional to the SpO₂ value of the blood.

An essential quality characteristic in comparing pulse oximeters is the resistance toward interferences. Filtering those unuseful signal portions which arise because of the movement of the patient is particularly problematic. Even with small movements, the amplitudes of the motion artifacts may seem larger than those of the pulse wave within the signal. If the signal is highly overlaid by motion artifacts, this will lead to a temporary operational failure of the equipment, with this problem being signaled accordingly. In the worst case, the equipment will not detect the distorted measurement and will not issue a signal, so that the measurement values indicated will erroneously be held as true. The quality of treatment of a patient may be clearly reduced due to measurement values being incorrectly indicated. Especially in the environment of operating rooms, the above-mentioned distortions represent a major disadvantage of pulse oximeters.

In addition to the motion artifacts, high-power light sources, such as those of operating-room lamps, fluorescent lamps or monitors, may cause unwanted interferences in the signal. With conventional pulse oximeters or plethysmographs, this problem is typically diminished by introducing additional measurement periods for determining the ambient light, and by subsequently subtracting the ambient-light measurement from the useful-signal measurement. During these measurement periods or time slots, all light sources of the sensor are switched off, and only the ambient light is measured. The ambient-light intensity is subtracted from the plethysmogram, and thus the portion of ambient light is largely separated from the pulse signal. However, especially with pulsating or AC-powered ambient-light sources, an interference portion will remain within the plethysmogram. The interference portion within the plethysmogram thus highly depends on the electronic equipment, or interferers, used in the surroundings. Especially in the intensive care of patients, a multitude of electronic devices and tools are employed, so that the susceptibility of pulse oximeters and plethysmographs to interference is a given fact particularly in intensive-care environments. Particularly in the field of intensive care, however, measurement errors of physiological parameters such as the heart rate or the blood oxygen saturation are extremely critical and may entail serious consequences.

In pulse oximetry, transmission and remission sensors have several LEDs (transmitters) and only one photodiode (receiver). The subject's tissue is irradiated by LEDs of different wavelengths, and the photodiode receives the light of different wavelengths from the tissue. In principle, it would be possible to differentiate various channels by means of the wavelengths of the LEDs, e.g. by color filters present at several photodiodes. Since this involves a large amount of technical expenditure on the side of the photodiode, the intensities of the LEDs may be modulated. Only then is it possible to differentiate between the wavelengths by means of a single broad-band photodiode.

In order to enable the receiver to differentiate between various transmit sources (LEDs) having different wavelengths, TDMA concepts (time division multiple access) are employed with known pulse oximeters. Each sensor LED has a time window assigned to it within which it is switched on. FIG. 25 illustrates this time sequence of signals. One may recognize that the various LEDs successively have time slots of equal durations associated with them which are separated by dark periods of equal durations. FIG. 25 shows a schematic sequence with three different LEDs. The LEDs of different wavelengths successively light up for a short time duration, in FIG. 25, the bright periods of the LEDs are designated by “LED 1”, “LED 2”, and “LED 3”. Typical frequencies with which the light sources of current pulse oximeters are controlled amount to 20-50 Hz. By adding additional dark phases during which none of the LEDs lights up, designated by “DARK” in FIG. 25, one tries to measure the signal portion caused by ambient light, and to subsequently subtract it from the useful signal. Nevertheless, the results are often distorted by ambient light or by high-frequency surgery influences. In high-frequency surgery, tissue is cut by means of high-frequency voltages. These high frequencies cause inductions in lines of the pulse oximeters and may thus interfere with their functioning. The local influences may be largely suppressed, since the sensors are protected against irradiation from the outside. Nevertheless, ambient light will enter into the shell of the sensor.

The signal quality is clearly improved by subtracting the ambient-light portion, determined by adding dark phases. However, interference artifacts will remain which may lead to incorrect SpO₂ values. Up to now, it has not been possible, despite numerous attempts, to remove those interferences, which are caused by fluorescent lamps, infrared heat lamps, operating-room illumination and monitors, from the useful signal. Since in pulse oximeters and plethysmographs, the ratio between useful signals, i.e. those signal portions caused by the change in volume of the tissue, and the interferences may be very unfavorable, those interferences which are distorted further by signal processing are also relevant. For example, prior to an analog/digital conversion, signals are low-pass filtered to avoid errors caused by subsampling. Since the filters used only ever have a finite attenuation within the stop band, errors caused by subsampling, also referred to aliasing errors, will nevertheless arise. Depending on the original interfering frequency, these interferences will then be mirrored into the useful range, where they may occur at different frequencies.

A further example of dynamic interferences may be found with subjects who have long-term measurements conducted on them. They wear a sensor with integrated LEDs and a photoreceiver over a relatively long time period for detecting long-term data. These patients or subjects, for example during car journeys through tree-lined streets or streets lined by many high buildings, are subject to pronounced and, as the case may be, rapid changes in the lighting conditions. In places, these changing lighting conditions express themselves in a manner very similar to the interferences in in-patient environments of hospitals. In principle, subjects subjected to long-term measurements are exposed to a multiplicity of ambient-light influences which may give rise to a whole spectrum of interferences.

The susceptibility of current pulse oximeters and plethysmographs to interferences will rise if the above-mentioned interferers are located within their surroundings. Especially in operating rooms or intensive-care units, there are a multiplicity of electronic devices, or electronic interferers. Particularly in such environments, thus, the susceptibility of current pulse oximeters and plethysmographs to interferences increases. This significant disadvantage may entail serious consequences for subjects if such situations give rise to measurement errors which cannot be immediately identified as such.

Known plethysmography methods may be found in the following documents, for example:

EP 1374764 A1/WO 2002054950 A08, which describes a basic circuit for measuring and detecting a plethysmogram, and deals with the above-described signal processing in detail.

EP 208201 A2/A3, wherein optical detection of a change of volume of a body part, and an evaluation device for evaluating the optical signals are protected, in principle. The method described there makes use of the changing outward volume change of extremities caused by the pulse and the changes in blood pressure associated therewith.

EP 341059 A3. Here, a basic pulse oximetry method is described which exploits light sources (LEDs) of different wavelengths. Light of different wavelengths is radiated through the subject's tissue, the light signals are absorbed from the tissue by means of optical sensors and are evaluated by a corresponding analog signal processing.

EP 314331 B1, a pulse oximetry method also based on light of different wavelengths is used for radiating the tissue of a subject. The optical signals thus obtained are converted to electric signals, and a value which provides insights into the oxygen saturation of the subject's blood is extracted therefrom.

EP 1254628 A1, the pulse oximeter protected here is also configured to determine oxygen saturation of blood, the method proposed here additionally reducing interferences caused by cross-talk.

U.S. Pat. No. 5,503,144/U.S. Pat. No. 6,714,803, here a description is given of signal processing methods for linear regression which determine an SpO₂ value by means of two plethysmograms. A correlation coefficient which serves as the reliability measure is determined from among the two plethysmograms.

DE 692 29 994 T2 discloses a signal processor taking up a first signal and a second signal correlated with the first signal. Both signals each have a useful signal portion and an unuseful signal portion. The signals may be taken up by the spreading of energy through a medium and by measuring an attenuated signal after transmission or reflection. Alternatively, the signals may be taken up by measuring energy created by the medium.

The first and second signals measured are processed so as to take up a noise reference signal which does not include the useful signal portions of the respective first and second signals measured. The remaining unuseful signal portions of the first and second signals measured are combined to shape a noise reference signal. This noise reference signal is correlated with each of the unuseful signal portions of the first and second signals measured.

The noise signal is then used to remove the unuseful signal portions within the first and second signals measured by means of an adaptive noise eliminating means. An adaptive noise eliminating means may be seen analogously to a dynamic multiband-stop filter which dynamically changes its transfer function in response to a noise reference signal and to the signal measured so as to eliminate frequencies from the signals measured which are also present in the noise reference signal. A typical adaptive noise eliminating means thus obtains the signal from which noise is to be eliminated, and a noise reference signal. The output of the adaptive noise eliminating means then is the useful signal with reduced noise.

US 2005/0187451 describes a method of use in a signal attenuation measurement for determining a physiological parameter of a patient. Further, a description is given of an apparatus for determining a physiological parameter of a patient from at least two signals which passed tissue of the patient and were attenuated there. In this context, the two signals are multiplexed using an FOCDM method (FOCDM=frequency orthogonal code division multiplex). The method enables separation of the two signals and suppression of external interference.

SUMMARY

According to an embodiment, an apparatus for reducing an interference portion in a time-discrete signal further including a useful portion may have: a first provider for providing the time-discrete signal including the interference portion and the useful portion, the first provider being adapted to sample an iterative optical signal which corresponds to bright periods during which a transmit light source adopts an on state; a second provider for providing a first time-discrete reference signal including a first frequency component of the interference portion, and a second time-discrete signal including a second frequency component of the interference portion, the first and second frequency components being shifted in phase, and the second provider for providing the first and second reference signals being adapted to sample two optical signals which correspond to dark periods during which no transmit light source adopts an on state; a subtractor for generating a differential signal from the two reference signals, the differential signal including a third frequency component caused by the first and second frequency components; and a manipulator for manipulating the time-discrete signal on the basis of the differential signal, such that in a manipulated time-discrete signal the interference portion is reduced.

According to another embodiment, a method of reducing an interference portion in a time-discrete signal further including a useful portion may have the steps of: providing a time-discrete signal including the interference portion and the useful portion, including sampling an iterative optical signal which corresponds to bright periods during which a transmit light source adopts an on state; providing a first time-discrete reference signal including a first frequency component of the interference portion; providing a second time-discrete reference signal including a second frequency component of the interference portion, the first and second frequency components being shifted in phase, and providing the first and second reference signals including sampling optical signals which correspond to dark periods during which no transmit light source adopts an on state; subtracting the two reference signals and providing a differential signal including a third frequency component caused by the first and second frequency components; and manipulating the time-discrete signal on the basis of the differential signal such that in a manipulated time-discrete signal the interference portion is reduced.

This object is achieved by an apparatus for reducing an interference portion in a time-discrete signal further including a useful portion, the apparatus comprising a first means for providing the time-discrete signal including the interference portion and the useful portion. The apparatus further comprises a second means for providing a first time-discrete reference signal including a first interference portion, and a second time-discrete reference signal including a second interference portion, the second interference portion being shifted in phase relative to the first interference portion. The apparatus further comprises a subtraction means for generating a differential signal from the two reference signals, the differential signal including a frequency component caused by the first and second interference portions, and a means for manipulating the time-discrete signal, on the basis of the differential signal, such that in a manipulated signal the frequency component is reduced.

In addition, the object is achieved by a method of reducing an interference portion in a time-discrete signal further including a useful portion, by providing the time-discrete signal including the useful portion and the interference portion, by providing a first time-discrete reference signal including a first interference portion, and by providing a second reference signal including a second interference portion, the second interference portion being shifted in phase relative to the first interference portion. In addition, the object is achieved by subtracting the two reference signals and providing a differential signal including a frequency component caused by the first and second interference portions, and by manipulating the time-discrete signal, on the basis of the differential signal, such that in a manipulated time-discrete signal the frequency component is reduced.

The core idea of the present invention is to reduce the interference portions which overlay the useful signal in plethysmography and pulse oximetry, in addition to subtraction of an ambient light measurement, by adaptive filtering. In pulse oximetry or plethysmography, interferences caused by sub-sampling (aliasing) occur in addition to the interferences caused by ambient light. The former interferences are mirrored from higher frequency ranges into the useful band and are shifted in phase in the individual channels due to the sub-sampling. By formation of a differential signal from the channels of the dark phases, a signal may be extracted which contains only those interferences which are caused by sub-sampling. On the basis of the interference portions in this signal, the interference portions in the bright phase channels may also be reduced.

BRIEF DESCRIPTION OF THE DRAWINGS

Embodiments of the present invention will be detailed subsequently referring to the appended drawings, in which:

FIG. 1 a) is a schematic representation of an inventive embodiment

FIG. 1 b) is a schematic representation of an inventive embodiment

FIG. 2 a) is a schematic representation of the irregular arrangement of the bright periods

FIG. 2 b) shows a regular arrangement of the bright periods in accordance with conventional pulse oximeters

FIG. 3 is block diagram of an implementation of the embodiment

FIG. 4 is a schematized representation of a spectrum of a signal within the baseband

FIG. 5 is a schematized representation of a spectrum of a signal within the transmission band

FIG. 6 is a schematic representation of two orthogonal chip sequences of a length of 101 chips

FIG. 7 is a schematic representation of a spectrum of a chip sequence of a length of 101 chips

FIG. 8 is a schematic representation of the signal within the transmission band

FIG. 9 is a schematic representation of the spread interference and de-spreading within the frequency range

FIG. 9 a) is a schematic representation of the spectrum within the baseband

FIG. 9 b) is a schematic representation of the spectrum of the chip sequence

FIG. 9 c) is a schematic representation of the spectrum within the transmission band

FIG. 9 d) is a schematic representation of the spectrum of the useful portions and interference portions within the baseband after de-spreading

FIG. 10 shows two exemplary received waveforms of two LEDs having different wavelengths

FIG. 11 is a representation of two exemplary waveforms for the dark duration and the ambient-light measurement, respectively

FIG. 12 depicts an exemplary transmission function of an extraction filter with an attenuation of 15 dB and a suppression of 100 Hz; enlargement in the area of 100 Hz

FIG. 13 depicts exemplary waveforms of the bright transmit channels from which the ambient-light signal has been subtracted, the enlargement shows the reference signal

FIG. 14 is a schematic representation of the block formation for further signal processing, l_(B) corresponds to the block length, l_(a) is a measure of the overlap

FIG. 15 a) shows an exemplary waveform of an input signal and of the low-pass filtered DC signal (DC portion)

FIG. 15 b) shows an exemplary waveform of the high-pass filtered signal (AC portion)

FIG. 16 depicts a model of the adaptive filter, with the input quantities on the left and the output quantities on the right, the reference signal being designated by W^(A) _(c)

FIG. 17 depicts an exemplary curve of a Kaiser-Bessel window having a block length of 256 points

FIG. 18 is an exemplary spectral curve of the normalized useful signals for the two bright transmit channels, red and infrared

FIG. 19 is an exemplary representation of the two spectra for red and infrared transmit channels, spectral values of identical frequencies being plotted against one another

FIG. 20 a) is a schematic representation of the least squares fit method for minimizing a vertical distance from a straight line

FIG. 20 b) is a schematic representation of the total least squares fit method for minimizing the actual distances from a straight line

FIG. 21 a) depicts an exemplary curve of the quotient between the red transmit channel and the infrared transmit channel at four different points in time k2

FIG. 21 b) shows an exemplary curve of a reference spectrum determined using the method of the complex total least squares fit method

FIG. 22 depicts an exemplary spectrum of a waveform, wherein the amplitudes of the interference are larger than the amplitudes of the pulse wave

FIG. 23 depicts an exemplary characteristic curve of a calibration function

FIG. 24 shows a basic block diagram of the hardware of a conventional pulse oximeter; and

FIG. 25 is a schematized representation of a time division multiple access method (TDMA).

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 a) shows a schematized representation of an inventive embodiment comprising an apparatus 100 for reducing an interference portion. A first means 110 for providing a time-discrete signal representing, for example, a disturbed signal of a plethysmograph. This signal may have come into being, for example, by radiating a tissue of a subject with infrared light. In addition, FIG. 1 a) shows a second means 120 for providing first and second reference signals representing, for example, two dark phase channels of a plethysmograph. These two channels initially contain interferences caused by influences of ambient light in the useful range, as well as interferences caused by sub-sampling. Since the interferences located within the useful range are in the same phase in the two reference channels, they may be masked out by formation of the differential signal. This is achieved by a subtraction means 130 for generating a differential signal from the reference signals. The differential signal is supplied to a means 140 for manipulating the time-discrete signal. The means 140 for manipulating obtains the time-discrete signal from the means 110 for providing the time-discrete signal, and manipulates it on the basis of the differential signal.

In a further embodiment, as is shown in FIG. 1 b), the apparatus 100 further comprises, in addition to the means already shown in FIG. 1 a), a means 150 for forming a weighted sum from the two reference signals, which is then supplied to the first means 110 for providing the time-discrete signal so as to reduce in-phase interferences in the time-discrete signal. In addition, the time-discrete signal manipulated by the means 140 for manipulating is supplied to a processing means 160, where the signal is processed further, for example a physiological parameter is extracted here.

In an inventive embodiment, a light source whose light is coupled into a part of a subject's body, and the signal received by a photodetector, is controlled such that it adopts, in a iterating sequence, the on state at irregular intervals. The irregularity causes an expansion to occur within the spectral range of the signal. The additional spectral components of the light signal give rise to additional interference immunity. In the simplest case, two spectral lines of the same level will arise. Since the probability that both spectral components will be interfered with at the same time is smaller than the probability that a single spectral component will be interfered with, a diversity gain arises in the frequency range. This diversity gain may be implemented by corresponding signal processing, so that increased interference immunity and, thus, increased reliability of the measurement of a physiological parameter is achieved by the irregular control of the light sources. In addition, a so-called spread gain results. By the irregular controlling, the energy of the useful signal is evenly distributed to several frequency portions. Since the irregularity is known, these energy portions may again be coherently overlaid within the receiver. Interference portions having the same frequencies are also overlaid within the receiver; however, since same are interdependent, a coherent overlay takes place here, so that a gain results for the useful signal. A narrow-band interferer which overlays the useful signal only at one frequency portion will experience, within the receiver, a spectral expansion which is analogous to that of the useful signal within the transmitter, since in both cases, signal portions are combined at irregular points in time.

This irregularity of the bright periods is depicted in FIG. 2 a in a schematized manner. FIG. 2 a shows an iterating sequence of a duration of ΔT. Within one sequence, a light source H₁ adopts an on state twice. This is indicated in FIG. 2 a by the entries H₁. During the other moments, when no entries are present within the time raster in FIG. 2 a, the light source is switched off. For comparison, FIG. 2 b depicts a sequence of a conventional pulse oximeter. FIG. 2 b shows a time division multiple access method (TDMA), wherein two light sources are controlled. During one sequence, each light source adopts the on state for one time slot. This is indicated by H₁ and H₂ in FIG. 2 b. During the other time periods, depicted by D₁ and D₂ in FIG. 2 b (D represents “DARK”), none of the two light sourced is to have adopted an on state.

Irregular controlling at the light source corresponds to a spread spectrum modulation. In combination with an inventive adaptive filtering connected downstream, the spread spectrum modulation reduces signals portions which are to be attributed to ambient-light influences or electromagnetic interfering sources (e.g. high-frequency surgery), and such signal portions which are to be attributed to the subsampling. In addition, subsequent signal processing also enables a particularly efficient measurement of the oxygen saturation in the blood and of the heart rate of a patient, it also being possible with the present method to perform reliable measurements even with a low arterial blood volume pulsation and while the patient is moving. The increased reliability of the measurement thus directly causes an increase in the quality of treatment for a patient. Thus, one advantage of the present invention is the fact that due to the increased reliability of the measurement values of a pulse oximeter, in particular in critical environments such as operating rooms or intensive-care units, increased chances of recovery and more efficient methods of treatment are enabled.

FIG. 3 shows an implementation of the embodiment. In FIG. 3, a spread spectrum modulation 300 is initially converted to an optical signal by an LED driver stage 305. In accordance with the spread spectrum modulation received, the LED driver means 305 couples light signals into a tissue 310 (e.g. into a finger), whereupon the light signals are modulated on their way through the tissue, and are subsequently received by a photoreceiver 315. Photoreceiver 315 converts the optical signals received to electrical signals and feeds same to an analog/digital conversion means 320 which converts the analog signal to a digital signal. The analog/digital conversion means 320 has a spread spectrum demodulator 325 connected downstream from it.

After the spread spectrum demodulation 325, the signal is adaptively filtered 330 in accordance with the invention and thereafter subject to a Fourier transformation 335. In a next step, a spectral mask 340 is now applied to the spectrum of the signal, whereupon the heart rate of the subject may be established and will then be output at an output 345. In a next analysis step, the so-called “complex total least squares fit” method 350, a variance of the difference of the different spectra which have been measured for light having different wavelengths may be determined, via a statistical analysis in the frequency range, and be output at the output 355 as a measure of reliability. Using the initial value provided by the “complex total least squares fit” means 350, an associated blood saturation value (SpO₂ value) may now be output, via a calibration function 360, at the output 365.

In order to be able to measure the light absorption of the tissue 310 with several light sources 305 having different wavelengths, and by means of a broad-band photoreceiver 315, a modulation method may be used, consisting of modulator 300 and demodulator 325. To better suppress interferences, the spread spectrum method is employed. This modulation method is based on the fact that because of the irregularity of the bright periods, the spectrum of the baseband signal is spread, or expanded. This effect is illustrated by FIGS. 4 to 9. Initially, FIG. 4 shows a spectrum |I(f)| of a baseband signal, the cutoff frequency of which is referred to as f_(B). With conventional modulation methods, such as amplitude modulation, the spectrum of the baseband signal is shifted into a frequency range more suitable for the transmission. FIG. 5 illustrates this case, and depicts the shifted spectrum |I_(A)(f)|. Such a spectrum results when the baseband signal is multiplied by a higher carrier frequency. The spectrum of the baseband signal remains unchanged in terms of its shape and energy. If this signal is overlaid by an interferer, it will not be possible to suppress this interference by demodulation, i.e. by means of a shift-back from the transmission band into the baseband. In the case of the spread spectrum modulation as is employed according to the invention, each transmit channel, which is understood to mean the transmit light signals of a wavelength, has a so-called chip sequence associated therewith which has been pre-computed. A chip sequence consists of a finite sequence of ones and zeros which are typically clocked in a frequency which is a hundred times higher than, by comparison, in a TDMA concept. The clock frequency is about 3 kHz. From a mathematical point of view, the chip sequences may meet certain characteristics in order to achieve the desired spreading action of the interfering signal, and to enable the reconstruction of the plethysmograms as well as of the ambient-light channels. In principle, the chip sequences may be orthogonal to be able to implement a channel separation in the demodulation, and so that a demodulation without any cross-talk is enabled.

FIG. 6 shows a schematic representation of two orthogonal chip sequences, the length of a chip sequence equaling 101 chips in the embodiment contemplated here. FIG. 6 depicts a time beam of a duration of 101 chip durations. The values of second chip sequences c(k) are plotted over these 101 chip durations. In the diagram, the two chip sequences are differentiated by dotted and solid lines, respectively. Whenever a chip sequence adopts a value of 1, this means that the light source associated is placed into the on state. It may be clearly seen in FIG. 6 that the two chip sequences are orthogonal, i.e. that the two associated light sources will never adopt the on state at the same time. In principle, it is also possible to employ chip sequences which simultaneously result in a 1, or it is possible to employ other sequences having other properties. However, particular emphasis shall be placed on the property of the sequences which causes the individual bright periods to occur at irregular intervals, so that a spectral spread is achieved. In addition, it may be clearly seen in FIG. 6 that the individual bright periods within a sequence are arranged in an irregular manner, and that there are points in time when both chip sequences take on the value of 0, i.e. when both light sources are switched off in the implementation.

A further important property of the chip sequences is that their spectrum should be spread as evenly as possible, so that the signal energy will spread as evenly as possible to a frequency range as broad as possible.

FIG. 7 shows the spectrum, i.e. the frequency range, of one of the chip sequences depicted in FIG. 6. It may be clearly seen in FIG. 7 that the spectrum of such a sequence is evenly spread, i.e. the spectrum is composed of equidistant identical values. The high direct component represented by the excessive value at the frequency of 0, may be explained in that the chip sequence can only adopt the values of 0 and 1. Thus, the sequence is not free from a mean value.

The spectrum of a chip sequence may therefore be regarded as a “comb” of equidistant carriers of identical amplitudes. The spectral equipartition of a chip sequence results in that a narrow-band interferer will be spread, after demodulation, into a broad-band noise. In the implementation of the advantageous embodiment is depicted in FIG. 3, the two LEDs are controlled using the chip sequences depicted in FIG. 6.

FIG. 8 shows the schematic representation of the signal of FIG. 4 within the transmission band |I_(c)(f). The baseband signal, as is depicted in FIG. 4, maintains its spectral shape, but its energy is distributed to many frequencies. This process is also referred to as spreading. If the signal depicted in FIG. 8 is interfered with by a narrow-band interferer, said interferer will be subject to spreading in the demodulation, whereas the energy portions of the signal of FIG. 8 will again overlay one another in a coherent manner within the baseband. Here, the demodulation corresponds to a renewed multiplication by the respective chip sequence. The result of the multiplication is then summed up across a length of chip sequences. Thus, if a receive signal is multiplied by one of the chip sequences, as are depicted in FIG. 6, it may easily be seen from FIG. 6 that by the multiplication, only those receive signals values are masked out from the receive signal which are received at a moment which correspond to a one in the respective chip sequence. These individual signal portions are then summed up across a chip sequence, as a result of which they will coherently, i.e. constructively, overlay one another. An interfering signal which has overlaid the receive signal is also masked in only at the respective moments in time. Also, the interfering signals are sampled at the respective moments in time and are summed up across the length of chip sequence. However, the interfering signals do not coherently overlay one another at the sampling moments, so that they will actually experience a spreading across the de-spreading, i.e. the multiplication by the chip sequence, so that after demodulation, these signals will be present in an attenuated form only.

FIGS. 9 a)-d) depict the operation of the spreading once again within the frequency range. FIG. 9 a) shows the spectrum of a signal within the baseband. FIG. 9 b) shows the spectrum of a chip sequence, the spectrum ideally being evenly distributed in spectral terms. FIG. 9 c) shows the spread baseband signal which now has energy portions at each individual frequency of the chip sequence. The energy of the baseband signal has been spread to the frequencies contained within the chip sequence. In the inventive implementation, the signal is received in this form from the tissue by the photosensor, the actual useful signal then was modulated to the spread signal through the tissue. FIG. 9 c) further depicts two interferences, “interference 1” and “interference 2”. The two interferences are narrow-band interferers as may be caused, e.g., by fluorescent lamps or high-frequency scalpels. FIG. 9 d) shows the spectrum of the signal after demodulation, or after de-spreading. It may be seen that the baseband signal has been reconstructed, and that additional frequencies of the interfering signals within the baseband have been added. FIG. 9 d) also shows that the remaining frequencies of the interference have clearly smaller amplitudes than the original interference itself, which is due to the spreading of the interfering signal.

Legendre sequences are chip sequences which meet the characteristics useful here and which exhibit good auto and cross-correlation characteristics. In the contemplated implementation of the embodiment, the sequences modulate two bright transmit channels and two dark transmit channels. The spectral properties of all sequences are identical and meet the equidistribution useful within the spectral range. In addition, a total of four sequences are considered, the four sequences being orthogonal to one another, which means that no two sequences will adopt the value of 1 at the same time. In principle, the use of other sequences is also feasible, the irregularity property of the bright periods is to be emphasized here, this does not presuppose that at any moment in time, only one sequence may have a bright period. Two of the four sequences are used in an implementation of the embodiment to control two LEDs having different wavelengths (red and infrared), the two remaining sequences serve to modulate ambient-light channels, they correspond to dark channels.

The LEDs are now controlled as monochromatic light sources via LED driver 305 of FIG. 3. The light of the LEDs which is modulated with the chip sequences passes through a tissue layer, and depending on the wavelength of the light source, it experiences an adequate attenuation in the process. The radiation of the LEDs, which is attenuated by the tissue, impinges at photoreceiver 315, is converted to a proportional photocurrent there, and is subsequently sampled, using an analog/digital converter 320, in a manner which is synchronous to the clock of modulator 300. The synchronicity between the modulator within the transmitter, and the AD converter and/or demodulator within the receiver may optionally be achieved by a control means which dictates clocks to both the transmitter and the receiver via control leads. The signal which has been synchronously sampled is fed to the spread spectrum demodulator 325. By means of the demodulation, spread spectrum demodulator 325 divides the signal of the photoreceiver up into individual channels. In a practically oriented implementation, these are two pulse channels for red and infrared LEDs, as well as two channels for measuring the ambient light. FIG. 10 shows two exemplary waveforms, the lower one corresponding to the red LED, and the upper one to the infrared LED. It may be seen in FIG. 10 that both signals are overlaid by a higher-frequency signal portion originating from the pulse signal of the subject, that both signals have a high direct component, and that both signals have a low-frequency pulsatile portion which may have been caused, for example, by changes in ambient light due to movements on the part of the subject.

FIG. 11 shows two exemplary waveforms for the two dark channels. In these two signals, too, it is possible to recognize the high-frequency portion originating from the pulse signal of the subject, as well as an interfering portion to be attributed to ambient-light changes. The direct component in FIG. 11 is correspondingly smaller than the direct component in FIG. 10, since both light sources are switched off during the dark channel phases. In order to specifically calculate, and extract, the influences of the ambient light from the bright transmit channels, the mean value of the two ambient-light channels is subtracted from the two bright transmit channels so as to remove that portion of ambient light which is below the two sampling frequencies from the signal measured. The formation of the mean value corresponds to the inventive formation of a weighted sum, as is implemented, in accordance with the invention, by a means 150 for forming a weighted sum in FIG. 1 b).

For demodulation purposes, a so-called matched filter is used, for each chip sequence, for extracting the transmit channels from the receive signal. Such a matched filter is an implementation of the spread spectrum modulator 325 of FIG. 3 and may be described as a mathematical operation with a chip sequence. The sensor signal is cyclically multiplied by the chip sequence, and the result is summed up over one chip-sequence length in each case. In the realization of the embodiment which is described here, these are the respective Legendre sequences. Mathematically speaking, the matched filter implements a scalar product between the chip sequence and the receive vector, i.e. the sampled receive signal. Transmitter and receiver are synchronized. The scalar product leads to blockwise de-spreading of a transmit channel into the baseband. What results at the same time is a subsampling with a factor which corresponds to the length of the chip sequence for the useful signal. To avoid aliasing, the bandwidth of the signal may be reduced prior to each subsampling operation. Thus, an anti-aliasing filter is useful which may be integrated, along with the matched filter, into one filter.

Studies have shown that interferences having large amplitudes are mainly due to artificial illumination. In Europe, the mains frequency is 50 Hz, the fundamental wave of the power (or of the intensity) is therefore 100 Hz, and its harmonic waves correspondingly amount to multiples of 100 Hz. Depending on the intensity of the interference, the attenuation of the extraction filter within the stop band is not sufficient. On the basis of these findings, the frequencies corresponding to a multiple of 100 Hz may be suppressed by adjusting the properties of the extraction filter (combined filter).

By way of example, FIG. 12 shows a transmission function of an extraction filter with an attenuation of 15 dB, wherein, additionally, the interferers are suppressed at multiples of 100 Hz. Thus, the extraction filter already includes a low-pass filter useful for subsampling, and at the same time a matched filter for de-spreading the spread signal from the transmission band into the baseband. A filter which implements subsampling is also referred to as a subsampler, the matched filter for de-spreading the spread signal is also referred to as a correlator, since it correlates a predefined chip sequence with the receive signal.

After extraction from the receive signal, the extracted and subsampled signals are present. The degree of subsampling is dependent on the length of the chip sequence. A sample of the useful signal results for each chip-sequence length due to the matched filter. By using several orthogonal chip sequences, several channels result during a chip-sequence duration, in the inventive embodiment, there are four channels, two bright transmit channels of the red and infrared LEDs, as well as two dark transmit channels, during which none of the transmit light sources adopts an on state, and which are used for ambient-light and interference compensation.

In addition, the interferences present above the useful band, i.e. interferences present above half the sampling frequency, are mirrored into the useful band at 15 dB by the extraction filter. The attenuation of the interferences above half the sampling frequency depends on the chip-sequence length. In the inventive realization of the embodiment, a chip-sequence length of 101 chips has been selected, which leads to an attenuation of 15 dB for interferences above half the sampling frequency. At the same time, the filter implements additional attenuation of all frequencies comprising a multiple of 100 Hz. FIG. 12 shows an exemplary transmission function of an extraction filter.

After the extraction filter, the useful signals are present within the baseband. To reduce the influence of the ambient light, in accordance with the invention, the ambient-light portion is subtracted from the useful signal, in accordance with the first means 110 for providing the time-discrete signal in FIG. 1 b). In addition, a differential signal is generated for the adaptive filter 330 in accordance with the second means 120 for providing the first and second time-discrete reference signals, and with subtracting means 130, as depicted in FIGS. 1 a) and 1 b). For ambient-light subtraction, a mean value is initially formed from the dark channels, which is then subtracted from the bright transmit channels. Depending on the type of chip sequences used, and/or depending on the configuration of the spectra of the individual chip sequences, it may be advantageous not to determine the exact mean value of the dark channels, but to linearly weight the dark channels. In the implementation of the inventive embodiment, Legendre sequences of the length of 101 chips are used. This implementation results in an optimal weighting of the dark channels of from 47.5% to 52.5%.

For the purposes of further inventive signal processing, it is important to differentiate between two frequency bands into which an interferer may be categorized. On the one hand, the band exists below half the sampling frequency, the useful band. On the other hand, the band exists above this frequency, the transmission band. Frequency components which are due to interference and are categorized into the useful band may be removed from the two useful signals (bright transmit channels of the red and infrared LEDs) by means of dark-phase subtraction. The signals of these frequencies are equal both in phase and in amplitude, and therefore do not appear in the difference of the two dark channels, the differential signal. An interferer within the useful band (or the baseband) will thus continuously result in 0 for the differential signal. An interferer within the useful band could be a light source which is detected by the photosensor through the tissue, and the intensity of which is modulated with the changes in volume of the arterial blood. However, these portions are not to be filtered out from the useful signal, as they contain the information desired (the pulsatile portion).

By contrast, an interferer could be categorized into the transmission band. In this case, the attenuation of the extraction filter will set in, which initially causes the interference to fall into the useful band in an attenuated state. In the implementation of the embodiment, this attenuation amounts to 15 dB. In addition, signals of these frequencies are subject to a phase shift which is different for each channel. This effect is to be attributed to the sequential sampling; although the orthogonal chip sequences are interlaced, cf. FIG. 6, they realize the samples at different moments in time. For signals above half the sampling frequency, this leads to a phase shift of the subsampled signals in the individual channels.

Thus, the difference between the two dark transmit channels (the differential signal) does not result in an extinction of these signals, but results in a signal, the frequency components of which contain the mirrored frequencies of the interferer from the transmission band. This signal now serves as a differential signal for an inventive adaptive filter 330, so as to reduce the remaining interferences from the transmission band as well. The ambient-light subtraction thus removes the interferences from the useful band, but also contains phase-shifted interfering portions from the transmission band. Once the interferences from the transmission band have experienced attenuation by the extraction, portions of this interference are now re-fed to the useful signal by the ambient-light subtraction. Therefore, what results is not the full attenuation for the interfering signals from the transmission band, but a smaller value. In the implementation of the inventive embodiment, the attenuation of the extraction filter initially amounts to 15 dB, but is reduced again by 3 dB by the ambient-light subtraction, so that a total attenuation of 12 dB results for interferers from the transmission band. FIG. 13 shows two exemplary waveforms for the two transmit channels, red and infrared LEDs, from which the ambient-light signal has been subtracted. In addition, FIG. 13 shows an exemplary differential signal in a magnified form.

For further signal processing, a block formation for the individual signals initially occurs. For this purpose, the signals are divided up into blocks of equal lengths, the individual blocks overlapping. FIG. 14 illustrates the block formation for further signal processing. Blocks of a length of l_(B) are formed from the samples of a useful signal, a new block being formed every l_(a) samples.

The useful signals are subsequently fed to a frequency-separating means. It is the task of the frequency-separating means to filter the direct component and the pulsatile portion from the input signals. In the implementation of the inventive embodiment, the separating frequency of the frequency-separating means amounts to about 0.5 Hz. FIG. 15 a shows the exemplary curve of an input signal being fed to the frequency-separating means. In addition, FIG. 15 a depicts the low-pass filtered portion (DC portion) of the input signal. FIG. 15 b) depicts the associated high-pass portion (AC portion) of the input signal. Further signal processing only relates to the high-pass portion of the input signal.

The high-pass filtered useful signals are now fed to an adaptive filter 330. It is the task of the filter, also referred to as interference canceller, to reduce interferences which were present in the transmission band and were mirrored, after demodulation, into the useful band in an attenuated state, cf. FIG. 9 d). The differential signal which contains the frequencies of the interference in the useful band has been formed by subtraction from the dark transmit channels. The differential signal differs in phase and amplitude from the interferences overlaid on the useful signals. The phase difference is caused by the sampling offset in time, the difference in amplitude is caused both by the sampling which is offset in time, and by the subtraction. It is therefore the task of the adaptive filter to filter out the undesired image frequencies from the useful signals using the differential signal. For this purpose, an interfering signal is constructed, from the differential signal, which is as close as possible to the interference overlaid upon the useful signal. There are several mathematical methods of determining the coefficients for adaptive filter 330. One known method would be to select the coefficients of adaptive filter 330 such that the deviation between the differential signal and the useful signal is minimized. For determining the coefficients, the complex total least squares fit method is also to be mentioned here.

FIG. 16 shows the model of the adaptive filter with the time-discrete input quantities {right arrow over (w)}^(A) _(r) and {right arrow over (w)}A_(i) for the two input signals of the bright transmit channels for red and infrared, A indicating that the input signals are high-pass filtered. In principle, the operations described below are applied to both input quantities separately, since the interference contemplated is present in them in a phase-shifted manner. The differential signal is also high-pass filtered as a matrix W_(c) ^(A), and forms the basis for determining the adaptive filter coefficients {right arrow over (λ)}_(r) and {right arrow over (λ)}_(i). Matrix W_(c) ^(A) results from blocks of the differential signal which are high-pass filtered to remove the direct component. The columns of the matrix each form a block of samples, e.g. of a length of 256. This block is indented in the matrix on a column-by-column basis, by one sample each, the matrix has as many columns as there are coefficients for the adaptive filter. The matrix may be described as

[W _(c) ^(A)]_(ij)(k)=w _(c) ^(A)(i+k·l _(a) +j)

∀jε{0, 1, . . . , N_(ifc)}

∀iε{0, 1, . . . , l_(B)−1},  (1)

wherein w_(c) ^(A) represents the high-pass filtered elements of the differential signal, k is a discrete control variable of the block formation, l_(a) is the jump constant in the block formation, l_(B) is the block length, and N_(ifc) is the filter arrangement, i.e. the number of filter coefficients of the adaptive filter, or interference canceller, reduced by one. The output of the adaptive filter thus is a weighted sum of blocks of the differential signals, the blocks having been shifted by one sample each. Using the adaptive filter, the interference vectors are initially reconstructed, which are designated by {right arrow over (w)}^(s) _(r) and {right arrow over (w)}^(S) _(i) in FIG. 16 and overlaid on input signals {right arrow over (w)}^(A) _(r) and {right arrow over (w)}^(A) _(i). By means of subtraction, the interference effects in the input signals {right arrow over (w)}^(A) _(r) and {right arrow over (w)}^(A) _(i) are reduced, as is depicted in FIG. 16.

What is initially sought for is a linear combination {right arrow over (λ)} of basis W_(c) ^(A) which best reproduces the input signal, i.e. a reconstruction of the interference has been overlaid on an input signal {right arrow over (w)}^(A).

{right arrow over (w)}^(A)=W_(c) ^(A){right arrow over (λ)}.  (2)

There are more equations than unknown variables, in that one assumes that the adaptive filter comprises fewer coefficients than the block length to be processed. This is why there is no specific solution in this case. What is sought for is a vector {right arrow over (λ)} which best fits the over-determined equation system:

∥W_(c) ^(A){right arrow over (λ)}−{right arrow over (w)}^(A)∥→Minimum.  (3)

This problem may be addressed using the pseudo inverse. Thus, using vector {right arrow over (λ)}, one obtains a linear combination of W_(c) ^(A) which may be used to describe the interference.

(W _(c) ^(A))^(#) {right arrow over (w)} ^(A)={right arrow over (λ)}.  (4)

Thus, the interferer may be reconstructed on the basis of the input signal:

{right arrow over (w)} ^(s) =W _(c) ^(A)(W _(c) ^(A))^(#) {right arrow over (w)} ^(A).  (5)

It may also be seen from FIG. 16 that the difference between the estimated interferer and the input signal results in the filtered signal:

{right arrow over (y)} ^(A) ={right arrow over (w)} ^(A) −{right arrow over (w)} ^(s),  (6)

or

{right arrow over (y)}={right arrow over (w)} ^(A) −W _(c) ^(A)(W _(c) ^(A))^(#) {right arrow over (w)} ^(A)(E−W _(c) ^(A)(W _(c) ^(A))^(#)),

wherein E represents an identity matrix. An alternative implementation of the present invention would be a filter which implements a notch filter on the basis of the knowledge of the frequencies occurring in the differential signal, which is switched into the path of the useful signal, and which attenuates the frequencies of the differential signal.

Since examinations are subsequently performed in the frequency range, the input signals are transformed into the frequency range by means of Fourier transformation. Due to the block formation, unwanted side effects result in the frequency range. A block formation is to be equated with a multiplication of a rectangular pulse, which masks out the very block under consideration from a receive signal, by the receive signal itself. If this block is subject to Fourier transformation, one will obtain, in the frequency range, a convolution of the Fourier-transformed rectangular pulse (since function) with the actual spectrum of the sequence of receive signal samples. To reduce the unfavorable effects caused by the convolution with the sinc function in the frequency range, the block of receive signal samples is multiplied, in the time domain, by a window function having a narrower spectrum than the sinc function. For implementing the embodiment, a Kaiser-Bessel function is used for this purpose. FIG. 17 depicts the waveform of a Kaiser-Bessel window by way of example. Multiplication of the signal blocks by the window function may optionally also be performed prior to the adaptive filtering.

For further signal processing, the two useful signals are now normalized. Subsequently, the Fourier transformation is conducted. After the Fourier transformation, the spectra may be presented in different views, such as their curve as a function of time or as a function of the frequency. FIG. 18 shows exemplary spectra of the normalized signals from the bright transmit channels red and infrared. The spectra show signals under good conditions, i.e. with relatively little interference. The Fourier transformation 335 is followed, in a next signal processing step, by applying a spectral mask 340 for determining the heart rate. The Fourier transformation of the two signals from the bright transmit channels initially provides two spectra. If the two signals were undisturbed, it would be possible to represent one of the two spectra as a linear combination of the other, respectively. However, since the two spectra comprise interferences, it is initially not possible to merge them into each other by a linear combination.

In FIG. 19, the two spectra for respectively identical frequency values are plotted against one another. One may see that the dots do not lie on a straight line, which would indicate a linear relationship between the two spectra. If the two spectra did not comprise any interferences, an original straight line would result with this representation. To solve this problem, an original straight line is now sought for using the method of the least squares fit, the sum of the square distances of all dots from this original straight line being minimized. This method is known by the synonym of the total least squares fit method.

FIG. 20 a) and FIG. 20 b) are to illustrate the approach in the total least squares fit method. Unlike the least squares fit method, which is depicted in FIG. 20 a), in the total least squares fit method the actual distance of a dot from a straight line is minimized, cf. FIG. 20 b). This approach initially leads to an over-determined equation system. The over-determined equation system may be solved using a singular value decomposition so as to find a solution which corresponds to the total least squares fit method. Using the singular value decomposition, initially the matrix representing the over-determined linear equation system is decomposed. This results in a matrix containing, on its diagonal, the singular values of the equation system. By maintaining the maximum singular value, and by setting all other singular values to zero, this matrix is reduced to rank 1, and the problem is thus traced back to a solvable linear equation system. Such a straight solution line is drawn in FIG. 19; it is located halfway between two other straight lines defining the range of values of valid slopes which result from the reference measurements of the SpO2 values. The slope of this straight line now represents a measure of the blood oxygen saturation of the subject. A reference spectrum may now be determined from the linear equation system which has been determined using the singular value decomposition.

The slope of the original straight line which has been determined in this manner may initially be distorted if a high-amplitude interference identically overlays in the two spectra. To reduce this type of interference, the spectral mask is employed. The function of the spectral mask 340 may be described as follows. In principle, it is a spectral method which browses the Fourier coefficients of the pulse signal within the spectrum so as to set all coefficients which do not belong to the pulse signal to zero. The principle of the spectral mask is based on the fact that the frequency components of the pulse wave differ from those of other interferers. The algorithm of the spectral mask fundamentally is a binary mask comprising the elements {0, 1}, by which the spectrum is multiplied on a point-by-point basis so as to suppress those Fourier coefficients which do not belong to the pulse signal.

FIG. 21 a) shows the exemplary curve of the quotient of two spectra of the waveforms of the bright transmit channels, FIG. 21 b) shows, in this context, the curve of a reference spectrum which has been corrected with regard to interferences. Both spectral curves are plotted at four different points in time, respectively, k2=1 . . . 4. If the quotient of the two spectra of FIG. 21 a) is compared to the reference spectrum across several time windows, it becomes clear that the quotient is correct only above the frequency components of the pulse signal, and is the same for all of these frequencies. Problem arise when the amplitudes of the interferences become larger than those of the pulse wave.

By way of example, FIG. 22 a shows a spectrum of a signals disturbed by interfering signals, the amplitudes of which are larger than the amplitudes of the actual pulse wave. The quotient of two spectra is undefined at the frequencies of an interferer, and it bears no relation to the blood oxygen saturation of a subject. Without the spectral mask, dominant features would lead to an incorrect blood oxygen saturation value, as is shown in FIG. 22. Studies have shown that such dominant interferences mostly occur within both spectra, i.e. within the spectrum of the red signal as well as within the spectrum of the infrared signal. This results in that, in the quotient formation, quotients of a value of 1 arise. A quotient of a value of 1 corresponds to a blood oxygen saturation value of approx. 80%. It is now the task of the spectral mask to differentiate the frequency components of the pulse wave from those of the interferers.

The spectral mask exhibits an algorithm of the harmonic relation. The method of the harmonic relation is based on findings of examining numerous pulse signals for their spectral properties. The fundamental finding is the harmonic relation of the three relevant frequencies f_(g) of the fundamental wave, f_(o1) and f_(o2) of the second harmonic. In this context, it is also known that the second harmonic is at double the frequency of the fundamental wave, and that the third harmonic is at three times the frequency of the fundamental wave. On the basis of this relation, a mask may now be created which masks in, in the frequency range, those frequency portions of double and three times the frequency of a fundamental wave, i.e. exhibits a 1 at these locations, and masks out all other frequencies, i.e. has a 0 at these locations. A sum may then be formed from the remaining coefficients, the sum being associated with the fundamental frequency. This process may then be repeated for any potential heart rates feasible, for example within a range from 30-300 Hz, and subsequently, that frequency at which the sum is maximized may be selected. A further property which may be taken into account in this context is that the amplitudes of the respective harmonics exhibit a decaying characteristic. This means that at the first harmonic or at double the frequency of the fundamental wave, the amplitude has a smaller amplitude than the fundamental wave itself. At the second harmonic, which has three times the frequency of the fundamental wave, the amplitude is smaller, in turn, than at the first harmonic. Values for which the appropriate condition of the decaying spectrum is not met will not be considered in the search for the maximum.

Now the heart rate may be determined via the position of the spectral mask. In the realization of the inventive embodiment of FIG. 3, the heart rate will be output at output 345.

After multiplication by the spectral mask, only the relevant frequency components were detected. According to the same principle as was already described, a quotient of the relevant spectra may now be determined again using the complex total least squares fit method and singular value decomposition. In this context, only those frequency components are used which have been determined by means of the spectral mask. Via these interference-corrected spectra, the original straight line and its slope may now be determined. In addition to the slope of the original straight line, it is also possible to extract a measure of the reliability of the slope determined from the matrix decomposition of the over-determined linear equation system. The variance in accordance with the Frobenius norm, which may be directly obtained from the matrix decomposition, indicates the similarity of the two signals.

The variance is used as an indictor of excessive interference effects which prevents the computation of the physiological parameters within the specified tolerance range. Then this variance can be output at output 355 in accordance with FIG. 3. The complex total least squares fit method has a calibration function 360 connected downstream from it. The slope of the original straight line which has been determined using the complex total least squares fit method and which is representative of the blood saturation value of the subject is passed on to a calibration function 360. The calibration function directly associates SpO₂ values (blood saturations values) with the slope values obtained. The respective SpO₂ values are then output, in accordance with FIG. 3, at output 365. FIG. 23 shows an exemplary characteristic curve of a calibration function. It can be seen how blood saturation values (SpO₂ values) are associated with quotients (ratio). The characteristic curves of the calibration function are empirically determined using reference measurements.

One advantage of the present invention is that adaptive filtering, which is specifically tailored to the field of application of plethysmography and pulse oximetry, and the combination of the specifically adapted adaptive filtering significantly improve the reliability of the plethysmograms and enable effective filtering of ambient-light interferences and interferences caused by electromagnetic fields (e.g. high-frequency surgery).

Another advantage is that using singular value decomposition for calculating SpO₂ values from the complex spectra, a measure of reliability in the form of a variance may be extracted and used for assessing the quality of the result, or that a malfunction may be reliably detected.

An additional advantage is that using the inventive apparatus for measuring the blood oxygen saturation and heart rate, reliable measurements may be made even at a low arterial blood volume pulsation while the patient is moving, which is due to the additional level of reliability obtained by the adaptive filtering.

In general terms, one may state that the quality of treatment for a patient, in particular in intensive care and in operating rooms, may be considerably improved by the present invention. Due to the increased reliability and robustness of the method, diagnostic errors due to distorted measurements and/or due to unreliable measurement values can be considerably reduced.

While this invention has been described in terms of several embodiments, there are alterations, permutations, and equivalents which fall within the scope of this invention. It should also be noted that there are many alternative ways of implementing the methods and compositions of the present invention. It is therefore intended that the following appended claims be interpreted as including all such alterations, permutations and equivalents as fall within the true spirit and scope of the present invention. 

1. An apparatus for reducing an interference portion in a time-discrete signal further comprising a useful portion, the apparatus comprising: a first provider for providing the time-discrete signal comprising the interference portion and the useful portion, the first provider being adapted to sample an iterative optical signal which corresponds to bright periods during which a transmit light source adopts an on state; a second provider for providing a first time-discrete reference signal comprising a first frequency component of the interference portion, and a second time-discrete signal comprising a second frequency component of the interference portion, the first and second frequency components being shifted in phase, and the second provider for providing the first and second reference signals being adapted to sample two optical signals which correspond to dark periods during which no transmit light source adopts an on state; a subtractor for generating a differential signal from the two reference signals, the differential signal comprising a third frequency component caused by the first and second frequency components; and a manipulator for manipulating the time-discrete signal on the basis of the differential signal, such that in a manipulated time-discrete signal the interference portion is reduced.
 2. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal and the second provider for providing the first and second time-discrete reference signals are implemented to provide the time-discrete signals by sampling analog signals using a sampler comprising a sampling frequency.
 3. The apparatus as claimed in claim 2, wherein the sampler is implemented to generate the time-discrete signal, the first and second time-discrete reference signals in that sampling is sequentially performed at different points in time at regular time intervals.
 4. The apparatus as claimed in claim 3, wherein the first provider for providing the time-discrete signal and the second provider for providing the first and second time-discrete reference signals each comprise a low pass whose cutoff frequency corresponds to at least the reciprocal value of half the sampling duration of the sampler.
 5. The apparatus as claimed in claim 4, wherein the manipulator for manipulating the time-discrete signal is adapted to manipulate frequency components in the differential signal in terms of their amplitudes and phase positions such that a difference between the differential signal and the time-discrete signal is reduced.
 6. The apparatus as claimed in claim 5, wherein the manipulator is implemented to filter the differential signal using a digital filter, the coefficients of the digital filter being set such that the difference between the time-discrete signal and the filtered differential signal is smaller than the difference between the time-discrete signal and the differential signal.
 7. The apparatus as claimed in claim 4, wherein the manipulator is adapted to filter the time-discrete signal using a digital filter and to set the coefficients of the digital filter such that frequency components in the time-discrete signal which come up in the differential signal are reduced in terms of their amplitudes.
 8. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal and the second provider for providing the time-discrete reference signals are adapted to sample optical signals.
 9. The apparatus as claimed in claim 1, further comprising a former for forming a weighted sum of the two time-discrete reference signals, which supplies the weighted sum of the two time-discrete reference signals to the first provider for providing the time-discrete signal, and the first provider for providing the time-discrete signal being implemented to subtract the weighted sum from an original signal and to provide the result as a time-discrete signal.
 10. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal is implemented to provide the signal on the basis of an infrared transmit light source or a red transmit light source.
 11. The apparatus as claimed in claim 1, wherein the first provider for providing a time-discrete signal is implemented to provide a signal which has passed a tissue of a living being.
 12. The apparatus as claimed in claim 1, wherein the apparatus for reducing an interference portion in a time-discrete signal is implemented to provide a signal comprising information on a physiological parameter.
 13. The apparatus as claimed in claim 1, wherein the apparatus for reducing an interference portion in a time-discrete signal is implemented to provide a signal comprising information on a heart rate or a blood oxygen saturation value of a living being.
 14. The apparatus as claimed in claim 1, wherein the first provider for providing a time-discrete signal is implemented to process an iterative optical signal, the optical signal comprising sequences, and a sequence comprising at least two bright periods during which a transmit light source adopts an ON state and at least one dark period during which no transmit light source adopts the ON state, and the at least two bright periods being irregularly arranged within a sequence, and the first provider for providing the time-discrete signal further being implemented to provide the time-discrete signal in accordance with a bright transmit channel on the basis of the information on the arrangement of the bright periods within the sequence.
 15. The apparatus as claimed in claim 1, wherein the second provider for providing the first and second time-discrete reference signals is implemented to receive iterative optical signals, the optical signal comprising sequences, and a sequence comprising at least two dark periods during which no transmit light source adopts an ON state and at least one bright period during which a transmit light source adopts an ON state, and the at least two dark periods being irregularly arranged within the sequence.
 16. The apparatus as claimed in claim 1, wherein a sequence of an optical signal is based on a clock in accordance with which the bright and dark periods occur.
 17. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal and the second provider for providing the first and second time-discrete reference signals are implemented to operate at a clock higher than 800 Hz.
 18. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal and the second provider for providing the first and second time-discrete reference signals are implemented to receive, to store or to generate a binary code word comprising a length and indicating the arrangement of the bright and dark periods, and to link the binary code word with an original signal in a block-by-block manner so as to acquire a digital signal of a transmit channel, the transmit channel being determined by the binary code word, and the block length being defined by the length of the code word, and a value of the transmit channel being extracted per block-by-block linking.
 19. The apparatus as claimed in claim 18, wherein the first provider for providing the time-discrete signal and the second provider for providing the first and second time-discrete reference signals are implemented to form a scalar product between the binary code word and a block of the original signal of the length of the binary code word, to weight the result, to link it with other results and to associate an overall result with a transmit channel.
 20. The apparatus as claimed in claim 1, wherein the sequence of the optical signal comprises at least two further bright periods originating from a second transmit light source, the first provider for providing the time-discrete signal being implemented to extract a further bright transmit channel using a further binary code word.
 21. The apparatus as claimed in claim 1, wherein the manipulator for manipulating the time-discrete signal is implemented to form columns of a matrix from the differential signal in a block-by-block manner, to regard these as coefficients of a linear over-determined equation system, and to solve same by an optimization criterion, as well as to adaptively track the coefficients.
 22. The apparatus as claimed in claim 1, wherein the first provider for providing the time-discrete signal is implemented to subdivide the time-discrete signal into blocks, multiply it by a window function, normalize it and/or determine its spectrum.
 23. The apparatus as claimed in claim 1, wherein the manipulated time-discrete signal is supplied to a processor implemented to determine a straight regression line in a large group of dots using the CTLSF method (complex total least squares fit) by means of a singular value decomposition.
 24. The apparatus as claimed in claim 23, wherein the processor is implemented to determine the linear coefficient using the CTLSF method (complex total least squares fit) for two linearly dependent spectra comprising independent interferences.
 25. The apparatus as claimed in claim 24, wherein the processor is implemented to determine the linear coefficient using a singular value decomposition.
 26. The apparatus as claimed in claim 1, implemented to determine, from the ratio of two spectral values of two bright transmit channels, blood oxygen saturation or an SpO2 value of an artery which has been radiated.
 27. The apparatus as claimed in claim 1, implemented to determine heart rate signals using a spectral analysis of bright transmit channels.
 28. The apparatus as claimed in claim 27, implemented to determine a spectral mask which insulates, in the frequency range, those signal portions from the spectrum which comprise the same spectral characteristic as a signal sought for.
 29. The apparatus as claimed in claim 28, implemented to determine a spectral mask for pulse signals of a living being with regard to a potential fundamental wave frequency and potential harmonic frequencies, as well as their relations to each other, from the spectra of the bright transmit channels.
 30. The apparatus as claimed in claim 29, implemented to infer a subject's heart rate from the relation of fundamental wave components and harmonic components within the spectrum of the bright transmit channels.
 31. The apparatus as claimed in claim 21, implemented to utilize, for spectral analysis, only such frequency portions of a spectrum which are relevant to a pulse portion.
 32. The apparatus as claimed in claim 1, wherein the processor is implemented to receive, to store or to generate a table which associates blood saturation values (SpO2) with quotients of spectral values.
 33. The apparatus as claimed in claim 1, wherein the processor is implemented to output a reliability measure of a physiological parameter determined.
 34. The apparatus as claimed in claim 23, wherein the processor is implemented to determine the reliability measure from the singular values of a matrix composed of two spectra of two bright transmit channels, a spectrum of the bright transmit channel representing a column of the matrix.
 35. The apparatus as claimed in claim 34, wherein the processor is to adapted to derive the reliability measure from the Frobenius norm of the matrix.
 36. The apparatus as claimed in claim 33, wherein the processor is implemented to determine a variance of the difference between two bright transmit channels or their spectra and to derive the reliability measure therefrom.
 37. The apparatus as claimed in claim 33, wherein the processor is implemented to determine a variance from the difference between bright transmit channels comprising reduced interference, or their spectra, and to derive a reliability measure therefrom.
 38. A method of reducing an interference portion in a time-discrete signal further comprising a useful portion, the method comprising: providing a time-discrete signal comprising the interference portion and the useful portion, comprising sampling an iterative optical signal which corresponds to bright periods during which a transmit light source adopts an on state; providing a first time-discrete reference signal comprising a first frequency component of the interference portion; providing a second time-discrete reference signal comprising a second frequency component of the interference portion, the first and second frequency components being shifted in phase, and providing the first and second reference signals comprising sampling optical signals which correspond to dark periods during which no transmit light source adopts an on state; subtracting the two reference signals and providing a differential signal comprising a third frequency component caused by the first and second frequency components; and manipulating the time-discrete signal on the basis of the differential signal such that in a manipulated time-discrete signal the interference portion is reduced. 